1. Field of the Invention
The invention relates to a technology of driving a stepping motor at low noise and low vibration.
2. Related Art
Hitherto, a stepping motor is used for applications in various position controls. A stepping motor is composed of a rotor and a stator having plural phases of windings and is arranged to rotate and stop by each unit angle. Control of the number of rotation steps allows the rotor to rotate or stop by a desired angle without feedback control. Such operational characteristic of the stepping motor is suited to position control application.
Recently, the stepping motor is used widely in adjustment of iris, focus or zoom as an optical system actuator in electronic imaging apparatus such as a digital still camera (DSC) or a digital video camera (DVC).
Operation of the stepping motor used in the digital video camera is particularly required to be low in noise and vibration. This is because noise generated by the stepping motor is captured by a built-in microphone to be recorded as noise, and vibration causes camera shake and lowered quality of recorded image. To meet such demand, driving technology of operating a stepping motor at low noise and low vibration is disclosed, for example, in patent document 1.
FIG. 15 is a block diagram of a conventional stepping motor driving apparatus disclosed in patent document 1. The diagram describes only constituent elements necessary for explaining the principle. Since the stepping motor has plural phases of winding and the construction is the same in each winding, only one phase of winding is shown.
The pulse width modulation controller 15 includes a comparator 16, a flip-flop 17, a reference pulse generator 18, and a conduction logic section 19. The reference pulse generator 18 sets the flip-flop 17 in every pulse width modulation period (PWM period). Hence the conduction logic section 19 turns on either one of transistors 6 and 9 and either one of transistors 7 and 8 which compose the switching section 5, in every specific period, in combination and timing so as not to shoot through. A current direction switch signal (PHASE in FIG. 15) entered into the conduction logic section 19 decides which one of transistors 6 and 9 and one of transistors 7 and 8 are turned on, and determines the direction of current flowing in the winding 3.
During turn-on of the transistors 6 to 9, electric power is supplied to the winding 3 from the power source 1, and the current flowing in the winding 3 increases. Hereinafter, the period in which the flip-flop 17 is set and electric power is supplied to the winding 3 with the increased current flowing in the winding 3 is called “PWM ON period”.
A supplied current measuring section 20 detects the current supplied in the winding 3 by turn-on of transistors 6 to 9 from the power source 1, and outputs the detected current value to a comparator 16. The supplied current measuring section 20 includes a detection resistor 21, sense amplifier 22, and gain setting resistors 23 and 24. An amplifier 25 includes a sense amplifier 22 and a gain setting resistors 23 and 24, and the amplification factor of the amplifier 25, that is, the gain from input to output of sense amplifier 22 is determined by the gain setting resistors 23 and 24. The current supplied to the winding 3 flows into the detection resistor 21, and the voltage generated across the detection resistor 21 is fed into the sense amplifier 22. The sense amplifier 22 multiplies the input voltage by the gain to send the multiplied voltage to the comparator 16 as a detected current value.
In the following explanation of operation, the current flowing in the winding 3 to be detected by the supplied current measuring section 20 is called “a detected current value”. The reference signal generator 14 generates stepwise waves increasing and decreasing in steps, and sends to the comparator 16 as a reference signal which indicates the current limit value. The reference signal expressing the current limit value generated by the reference signal generator 14 is a current target value for the winding 3.
The comparator 16 compares the entered detected current value with the current target value, and resets the flip-flop 17 when the detected current value exceeds the current target value. By resetting the flip-flop 17, the conduction logic section 19 turns off both transistors 7 and 8 for composing the switching section 5. While the flip-flop 17 is reset and transistors 7 and 8 are turned off, power supply from power source 1 to winding 3 is cut off, and the current flowing in the winding 3 is decreased by regenerative operation.
While both transistors 7 and 8 are turned off, if both transistors 6 and 9 are cut off, the current flowing in the winding 3 is regenerated by either one of the flywheel diodes 11 and 12, and either one of the flywheel diodes 10 and 13. While both transistors 7 and 8 are turned off, if both transistors 6 and 9 are turned on, the current flowing in the winding 3 is regenerated by transistors 6 and 9.
While both transistors 7 and 8 are turned off with either one of transistors 6 and 9 turned on, if the flywheel diode connected to the transistor not turned on is at forward bias, the regeneration current is caused by either one of flywheel diodes 11 and 12, and either one of transistors 6 and 9. If the flywheel diode connected to the transistor not turned on is at backward bias, the current regeneration is caused by either one of flywheel diodes 10 and 13, and either one of transistors 6 and 9.
A period for which the flip-flop 17 is reset and the current flowing in the winding 3 is decreasing by the regenerative operation is called “PWM OFF period”. During PWM OFF period, the current flowing in the winding 3 decreases. However when the output signal of the reference pulse generator 18 sets the flip-flop 17 again, it is changed to PWM ON period, and the current flowing in the winding 3 begins to increase again.
By this operation, the average current supplied to the winding 3 gradually approaches the current target value. As the current target value increases or decreases stepwise, the average current supplied to the winding 3 increases or decreases stepwise, and the operation is the same in other phases of windings than winding 3, and therefore the stepping motor 2 rotates and operates at rotating speed depending on the speed of step advancing.
The current target value generated by the reference signal generator 14 is described. FIG. 16 is a diagram showing the relation of a reference signal and a current direction switch signal in a conventional stepping motor driving apparatus.
The reference signal generator 14 generates a stepwise wave which increases and decreases in steps, sends it to the comparator 16 as a current target value. As the current target value increase or decreases in steps, the stepping motor rotates by each unit angle. Step advance of the current target value is determined by input of CLK (clock signal) instructing the step advance, but it can be also determined by counting of step advance interval by a timer. The step advance period of the current target value is determined by input CLK period or period of a timer for determining the step advance interval. The period for advancing the step of the current target value determines the period of the stepping motor for rotating a unit angle is determined, and further the rotation period of the stepping motor is determined. The current target value is preferred to be a sinusoidal signal in terms of low noise and low vibration. The reference signal generator 14 generates a stepwise wave by sampling a sinusoidal wave.
FIG. 16 shows a stepwise wave sampled in 64 steps as a current target value. Along with advance in steps, each value of the stepwise wave obtained by sampling the sinusoidal wave at each step is outputted sequentially, resulting in the stepwise wave sampling the sinusoidal wave.
Current direction of a current flowing in the winding 3 is specified by a current direction switch signal as shown in FIG. 16. That is, each value of the stepwise wave shows the amount of the current target value, and the current direction switch signal shows the direction of current. Further, to avoid sudden current changes due to stepwise level change, the stepwise wave smoothed by integrating means such as low pass filter is sent to the comparator 16 as a current target value.
The stepwise wave sampling a sinusoidal wave is not always required. In terms of mounting area, a stepwise wave sampling pseudo-sinusoidal wave, or stepwise wave out of sinusoidal waves may be also used. If sudden current changes by stepwise level changes may be permitted, unsmoothed stepwise waves may be sent to the comparator 16.
* * * Patent Document 1: JP-A-2004-215385
According to the conventional steeping motor driving apparatus, however, waveform of a current flowing in the winding 3 maybe distorted due to the response delay of the sense amplifier 22.
This problem is discussed by referring to FIG. 17 to FIG. 22B.
FIG. 17 is a circuit diagram of general sense amplifier structure and PWM OFF period operation point. The sense amplifier 22 include P channel MOS transistors 30a, 30b and 30c, N channel MOS transistors 31a, 31b and 31c, and differential transistors 32a and 32b, a current source 33, and a phase compensation capacitor 34. The gain setting resistors 23 and 24 have the same resistance value R, with the gain doubled.
FIG. 18 shows general sense amplifier structure and PWM ON period operation point. FIGS. 19A to 19C are current path diagrams when changing the phases (reference sign “35” in the diagram denotes a current path). FIGS. 20A to 20E are current waveform diagrams when the current target value is large in the conventional stepping motor driving apparatus. FIGS. 21A to 21E are current waveform diagrams when the current target value is small in the conventional stepping motor driving apparatus. FIG. 22A and 22B are waveform diagrams showing current waveform distortion in the conventional stepping motor driving apparatus.
During PWM OFF period, because of regenerative operation explained above, a current does not flow in the detection resistor 21. As a result, a grounding voltage is supplied to the non-inverting input terminal of the sense amplifier 22, as shown Vin+=0 V in FIG. 17.
The sense amplifier 22 cannot output a voltage lower than the minimum voltage determined by a constant current flowing from P channel MOS transistor 30c and ON resistance of N channel MOS transistor 31c. Even if an amplifier of so-called rail-to-rail type is used, 0 V cannot be outputted when the minimum voltage of the sense amplifier 22 is 0 V.
In FIG. 17, the minimum voltage is 20 mV, and Vout is 0.02 V. At this time, a half voltage, that is, 10 mV is fed to the non-inverting input terminal of the sense amplifier 22 owing to its structure, showing in FIG. 17 as Vin−=0.01 V. In the state shown in FIG. 17, relation of virtual grounding of the sense amplifier 22 is broken, and differential transistors 32a and 32b are not in balanced state, and a voltage nearly equal to the voltage of power source 1 is applied to the phase compensation capacitor 34. FIG. 17 shows it as Vc=VCC.
Hereinafter, the state in which relation of virtual grounding is broken is called that the loop of the sense amplifier is out. An electric charge of [Ccomp×(VCC−20 mV)] is accumulated in the phase compensation capacitor 34, where Ccomp is the capacitance of the phase compensation capacitor 34.
FIG. 18 shows an operation point of the sense amplifier during PWM ON period. In PWM ON period, since a current flows in the detection resistor 21, a voltage determined by the current flowing in detection resistor 21 and resistance of the detection resistor 21 is applied to the non-inverting input terminal of the sense amplifier 22. In FIG. 18, it is shown as Vin+=0.2 V. At inverting terminal of the sense amplifier 22, 0.2 V is fed, and the sense amplifier 22 outputs 0.4 V.
In FIG. 18, Vin− is 0.2 V and Vout is 0.4 V. At this time, the relation of virtual grounding of the sense amplifier 22 is maintained, and the differential amplifiers 32a and 32b are in balanced state. A gate voltage Vgs1 of N channel MOS transistor 31c is applied to the phase compensation capacitor 34, so that the voltage determined by a constant current flowing from P channel MOS transistor 30c and ON resistance of N channel MOS transistor 31c may be 0.4 V. FIG. 18 shows it as Vc=Vgs1.
Hereinafter, the state in which the relation of the virtual grounding is maintained is called that the loop of the sense amplifier is maintained. An electric charge of [Ccomp×(Vgs1−0.4 V)] is accumulated in the phase compensation capacitor 34.
Even when a voltage is supplied from the detection resistor 21, if the loop of the sense amplifier is out, the sense amplifier 22 does not respond and the detected current value cannot be judged correctly. To judge the detected current value correctly after transition from PWM OFF period to PWM ON period, it is required to transfer from the operation point shown in FIG. 17 to that shown in FIG. 18, and in particular, the electric charge in the phase compensation capacitor 34 is a problem.
As mentioned above, an electric charge of [Ccomp×(VCC−20 mV)] is accumulated at the operation point shown in FIG. 17, and an electric charge of [Ccomp×(Vgs1−0.4 V)] is accumulated at the operation point in FIG. 18. The detected current value cannot be judged correctly unless an electric charge of [Ccomp×4.4 V] is discharged, where VCC=5.02 V and Vgs1=1.0 V. The time required for discharge is the time until the sense amplifier 22 can correctly judge the detected current value after transition from PWM OFF period to PWM ON period, and it becomes hence “a detection delay”.
Such discharge is caused by a difference in currents flowing in the N channel MOS transistor 31b and the differential transistor 32b. As the differential transistor 32b is turned off more completely (as the larger voltage is input to the non-inverting terminal of the sense amplifier 22 after transition to PWM ON period, the differential transistor 32b is turned off more completely), the required discharge time becomes shorter, and the detection delay is reduced.
To the contrary, as the input voltage to the non-inverting terminal of sense amplifier 22 is smaller after the transition to PWM ON period, that is, as the current flowing in the detection resistor 21 is smaller, the differential transistor 32b is turned off more poorly, and the required discharge time becomes longer, with the detection delay being longer. Therefore, the detection delay appears more significantly at driving steps of the smaller current target value, such as driving steps=0, 31 to 33, 63 shown in FIG. 16. Since the driving step of the small current target value is close to the point of inverting the polarity of current, this is called “zero cross” hereinafter. In FIG. 16, the zero cross is indicated by point A.
In PWM OFF period explained above, the loop of the sense amplifier is out, and the loop of the sense amplifier may not be out also in PWM ON period.
The operation when the loop of the sense amplifier is out in PWM ON period is explained by referring to FIG. 16 and FIGS. 19A to 19C.
In PWM ON period, as shown in FIG. 19A, power is supplied to the winding 3, and a current flows into the supplied current measuring section 20. In FIG. 19A, transistors 8 and 9 turn on, and transistors 6 and 7 turn off. In PWM OFF period, on the other hand, because of the regenerative operation as shown in FIG. 19B, a current does not flow into the supplied current measuring section 20. In FIG. 19B, the transistor 9 turn on, and the transistors 6, 7 and 8 turn off. In PWM OFF period shown in FIG. 19B, the current flowing in the winding 3 attenuates. But at driving step=32 or 0 shown in FIG. 16, since the current is small, the voltage applied across the winding 3 is small in PWM OFF period, and hence the current flowing in the winding 3 hardly deteriorates.
When the advancing time of driving steps is short, that is, when the rotating speed of the stepping motor is fast, the current of winding 3 does not attenuate fully to 0 in transition to the next driving step. When the driving step transits from 32 to 33 or from 0 to 1 with the current left over in the winding 3, the current direction switch signal is changed over and the current at the winding 3 is inverted. Hence, transistors different from that in one driving step before turn on, as shown in FIG. 19C. In FIG. 19C, the transistors 8 and 9 turn off, and the transistors 6 and 7 turn on. At this time, the current at the winding 3 flows from the ground to the power source, and the current flows into the supplied current measuring section 22 reversely from the ground, and the current further flows into the detection resistor 21 reversely from the ground.
As a result, a negative potential is generated across the detection resistor 21, and is also applied in the sense amplifier 22. When the negative potential is applied, the loop of the sense amplifier is out with the same reason as in the case of input of grounding potential mentioned above, and the detection delay occurs. Therefore, as indicated by A in FIG. 16, right after changeover of the current direction switch signal PHASE, that is, immediately before inversion of a current of the winding 3, the loop of the sense amplifier is out even after the transition to PWM ON period, and the detection delay occurs.
Current waveform in the case of the detection delay is explained by referring to FIG. 20 and FIG. 21. In FIG. 20 and FIG. 21, the portion indicated by A is the detection delay. In FIG. 20, during the detection delay, the detected current value does not exceed the current target value. In this case, if there is a detection error, there is no adverse effect on detection operation.
When the attenuation in PWM OFF period is large, it takes a long time until reaching the current target value after the transition to PWM ON period. Thus the actual current does not reach the current target value during the detection delay, and it is highly possible that adverse effect does not occur as shown in FIG. 20. The higher the current target value, the larger is the attenuation in the regenerative operation in PWM OFF period, and at driving step of high current target value, the effect is none or very small.
In FIG. 21, during the detection delay, the detected current value is over the current target value. In this case, since the detection is not conducted during the detection delay, although the current exceeds the current target value, the PWM ON period continues, and hence it is out of the current target value. When the attenuation in PWM OFF period is small, it takes only a short time to reach the current target value after the transition to PWM ON period. Thus the actual current reaches the current target value within the detection delay, and hence it is highly possible that adverse effects occur as shown in FIG. 21.
The lower the current target value, the smaller is the attenuation in the regenerative operation in PWM OFF period, and it is highly possible that adverse effects occur at driving step of the low current target value. It means, particularly near zero cross, that the waveform is distorted obviously due to deviation from the target current. That is, as shown in portion A in FIG. 22, near zero cross, the current is deviated to the larger side from the current target value, and the waveform is distorted.
Thus, according to the conventional stepping motor driving apparatus, due to the detection delay of the sense amplifier, obvious distortion of the waveform may occur near zero cross in particular. Due to the waveform distortion, vibration and noise cannot be decreased sufficiently in application, more particularly, to an electronic imaging apparatus, and there is a further demand for lower vibration and lower noise of the stepping motor operation.
The invention is directed to the above problems, and hence has an object to present a stepping motor driving apparatus and method capable of lowering vibration and noise in operation of the stepping motor.